Tuesday, December 22, 2015

Troubleshooting EMI Seminar with Lee Hill: Interview

By Senior Technical Editor, Kenneth Wyatt.

This interview was conducted on Dec 9th, 2015 in anticipation of the upcoming Troubleshooting EMI with Lee Hill: Identify, Characterize, and Prevent Interference Problems Seminar Tour hosted by Rohde & Schwarz and Interference Technology.

The tour is running from March 10-24, 2016 in five different cities - Austin, TX; Irvine, CA; Milpitas, CA; Livonia, MI; and Chelmsford, MA. Attendance for the seminar is free, and engineers can reserve a place now by signing up online HERE.

Contact emily.webster@rsa.rohde-schwarz.com with any questions.

Kenneth Wyatt: What changes in technology are driving today’s EMC issues?

Lee Hill: One observation is that it’s easier for cheaper, smaller things to run fast. The other is that more products are networked or incorporating wireless technologies. An example is home exercise equipment – in the past these devices were purely mechanical. Today, exercise equipment you bring into your home is now loaded with electronics, variable speed motor drives, networking capability, and even video and audio entertainment. Many low-end appliances are full of high-speed stuff.

There’s also such a high-level integration of silicon. Single ICs can contain high-speed video, Wi-Fi, RFID, etc., all in the silicon. Other products I’ve worked on include drones and all the remote control aspects of real-time flight. Wireless self-interference (digital noise reducing on-board receiver sensitivity) has also been more of an issue.

There’s been a revolution in silicon design and simulation for EMC outside the U.S. – especially in vehicular design. I think the silicon has gotten so much better that system-level radiated emissions, radiated immunity, and ESD have improved dramatically over the years.

KW: What do you find are the top product design issues that are experienced?

LH: The top design issues really depend on who you’re talking to. IC manufacturers might say a different set of problems is their main EMC nemesis.

In the area of EMC, the common problem is a lack of understanding of EMC engineering fundamentals – for example, identifying dipole antennas (antenna structures that can radiate) and coupling paths. Generally, engineers can identify the source and victim, but they have no clue where to go from there.

KW: About the Troubleshooting EMI Seminar with Lee Hill sponsored by Rohde & Schwarz, I think it’s great you’ll be presenting in so many cities around the U.S. It seems you’ll be hitting some of the top tech areas on both coasts and in the Midwest.

LH: Looking forward to it! (Tour dates will take place in Austin, TX; Irvine, CA; Milpitas, CA; Livonia, MI; and Chelmsford, MA.)

KW: Why is Rohde & Schwarz doing this now?

LH: That’s a great question, and I have asked Faride Akretch, Segment Marketing Manager of Rohde & Schwarz, to respond:

Faride Akretch: We want to provide value to our customers. And besides compelling products, we also deliver educational content and activities on a regular basis. Those can be application notes, white papers or customer trainings like this one. Sometimes these seminars are being held in person, or virtually like our sponsorship and participation in EMC Live hosted by Interference Technology. We’ve identified a real need for an increased focus on EMC and EMC pre-compliance. It seems that with so much integration, EMC failures have become more difficult to understand and that customers want and need to dig deeper as to where possible interferers are originating from.

KW: One more question for Faride Akretch. Are there changes in measurement technology that is driving this Troubleshooting EMI Seminar with Lee Hill?

FA: As part of the Seminar, we will be highlighting some newer real-time tools that allow visualization of the spectrum yielding a very powerful EMI diagnostic capability. We’ll also be including probing solutions and how to approach testing and EMC pre-compliance tests.

KW: And now Lee Hill, while many of the EMC compliance tests continue to specify a standard EMI receiver or swept spectrum analyzer as dictated by the standards, real time analysis has become much more affordable. Will you be incorporating some of this newer measurement technology in your presentations?

LH: Yes, we’ll be demonstrating how to use real-time spectrum analysis to help identify issues that otherwise would be completely hidden.

KW: So, why should product designers attend these seminars? Why is this subject important?

LH: Some of the important takeaways would include discovering tools they were unaware that existed, learn why these tools are so incredible and useful, learn how to use them, and to have fun watching them be used.

KW: Finally, I know your courses include lots of real-time demonstrations. Will you be able to do the same for this seminar?

LH: The seminar will contain live demonstrations (not prerecorded video) of EMC measurements and troubleshooting ideas using measurement tools and real-life products that are familiar to EMC engineers and representative of commonly encountered noise problems in industry.

Registration for this free seminar is open now. For more information and to register for a date, CLICK HERE.

Lee Hill is an industry expert in electromagnetic compatibility and founding partner of SILENT Solutions LLC, an EMC and RF design firm established in 1992. Lee provides EMC troubleshooting services, design reviews, and training to a wide variety of industries nationally and around the world and also is a member of the adjunct faculty at Worcester Polytechnic Institute (WPI) where he teaches graduate-level classes in EMC. Lee also teaches at the University of Oxford (England), and for the IEEE EMC Society’s annual Global University and EMC Fundamentals program. He earned his MSEE in electromagnetics from the Missouri University of Science and Technology EMC Laboratory.

Thursday, July 9, 2015

The Simplified Method and the Near-Field Catastrophe

The simplified method, and in particular the system base-RF-power determination used in the simplified method equation, only applies to far-field data. Why? Well, as an example, the radiation polar-plot provided by antenna suppliers is far-field data and in no way represents how the field-strength varies with angle close to the antenna.

So where exactly is the near-field / far-field divide? That is, where does the simplified method no longer apply? We could talk about this until the cows come home and get nowhere, because basically one man’s near-field is another man’s far-field, with both parties eager to prove how the dividing line was derived quoting various permutations of wavelength and Pi as the precise answer.

The Near-Field / Far-Field Decree

So for the purposes of this blog-thread we are going to create a dictate. As regards RF waves surrounding a radiating antenna, when a wave is an independent entity the wave is in the far-field. By independent we mean the excitation signal feeding the source can do a backward-summersault for all the wave cares, it has already escaped the influence of the source and will carry on along it’s merry way at the speed of light (in vacuo).

Conversely, if a change in the excitation signal causes a change in a surrounding RF wave, that wave is in the near-field. For example, if the excitation source to a resonating dipole was suddenly removed the voltage and current standing waves present across the resonating structure would die away almost instantaneously.

We complete the dictate by stating that the near-field / far-field dividing line is at 3 wavelengths.

Effect on the Simplified Method

The new decree means the lowest frequency at which the simplified method is valid is 300MHz. The determination is easy enough:-

Our test distance is 3 meters. Obeying the decree means that in order for the calibration plane to just be in the far-field, the 3 meter distance must represent 3 wavelengths. Therefore the limiting wavelength is 1 meter. Using c = f/Lambda, this corresponds to a lowest frequency of use of 300MHz.

The Antenna Suppliers’ Get-Around

Given antenna suppliers sell antennas starting at 80MHz (wavelength 3.75 meters) into this 3m RF immunity test market, how do they get around that the supplied gain and radiation pattern data only applies at far-field distances? (This of course is where the “yes it is, no it isn’t in the near-field” brigade kick in and try to claim that 3m distance is far-field even at 80MHz. And this of course is why we ‘cut them off at the pass’ by announcing the decree).

The supplier get-around is to provide data on how much RF power is required to produce a particular test-field at a particular test-distance. The antenna supplier is not forthcoming with (or doesn’t know) how much of the test-field is created by field type, that is how much of the measured field is contributed by the far-field and how much by the near-field. In fact you will find that the only data supplied is boresight data. That is the data represents the field we would measure with a field-probe mounted at the center of the calibration plane. No guarantee is given regarding the field achieved elsewhere across the calibration plane. Why does this matter? Answer – we are designing the test system under ideal conditions (perfect, very large fully-anechoic chamber). The premise being that if we are unable to generate a compliant field across the calibration plane under these ideal conditions, we do not stand ‘a snowflake’s chance in hell’ of achieving one under real 3 meter semi-anechoic chamber conditions. For instance, what if the measured field across the plane was to vary due to multiple peaks and nulls in the near-field radiation pattern?

To be continued...

Tom Mullineaux
Lionheart Southwest

Friday, June 5, 2015

Incorporating Antenna VSWR Loss into the Simplified Method

The Simplified Method of Establishing the RF Power Required by a RF Immunity System Continued...

Previously we presented the novel idea that the RF power required by a RF immunity system is a base RF power level (determined entirely by constants), minus the gain of the antenna. The basic principle is shown pictorially in Figure 1. The picture says the system RF power requirement in dBm equals the base RF power level in dBm, minus the antenna gain in dBi. A quick sanity check says this makes sense in that the higher the antenna gain, the less power required by the system.

Figure 1

The approach is also self-correcting in that should the linear gain of the antenna drop to less than 1 as can happen in lower frequency test systems (say SANT / SISO = 0.9), the gain in dBi will become minus, resulting in the system power requirement equaling the base power level PLUS the antenna gain.

So far so good.

We then built on this by converting all phenomena requiring more system RF power into loss-blocks and adding the loss blocks to the system diagram as shown in Figure 2.

Figure 2

Later on, we will simply add the overall dB loss of the blocks to the antenna dB gain to obtain the ‘overall system gain’ such that we are back to Figure 1, with the rightmost block amended to represent the system gain.

This approach, combined with the graphical representation described in the AH Systems webinar [Link Here] provides superb understanding / visualization of the system behavior across the band of interest, and uses the power of dB notation to simplify power computation.

In this particular blog entry we will concentrate on the mismatch presented to the system by the antenna.

Generating the VSWR Loss-Block

We need to convert the antenna VSWR phenomenon into the ‘basic loss-block form’ shown in Figure 3.

Figure 3

Please Note: this section also gives the solution to the teaser question posted last time.

The VSWR loss-block must obey the form of the equation in Figure 3, that is:

We need to establish Pout / Pin. We start with the classic VSWR representation as applied to an antenna shown in Figure 4a. Note in all cases below, the antenna symbol represents any type of antenna and the measurement plane is at the antenna connector.

Figure 4a

The figure shows the incident voltage Vinc striking the measurement plane, a portion of Vinc passing through the measurement plane to the antenna (shown as Vnet, the net voltage used by the antenna to create the test field), and a portion of Vinc being reflected back along the transmission line (the reflected voltage Vref).

By inspection we can see that Vinc is the input to our loss block and Vnet is the output. Again by inspection

Here we introduce ρ, the reflection coefficient at the measurement plane. This is equal to the ratio Vref / Vinc, so Vref can be written as ρVinc. This is shown in Figure 4b.

Figure 4b

Figure 5

We still want Pout / Pin so we need to convert Figure 4a to the power based diagram of Figure 5

So the

of Figure 4a becomes the

of Figure 5

To do this we convert all voltages to power with reference to the characteristic impedance of the transmission line (cable). That is: -

For future reference that is


So the

Of Figure 5 becomes

Pnet (that is Vnet2 / Zo ) is the output of our VSWR loss-block in Figure 2 and Pinc (Vinc2 / Zo) is the input

Rearranging to make Vnet2 / Zo the subject of the equation

But looking again at Figure 4b, Vref = ρVinc, where ρ is the reflection coefficient presented at the measurement plane

That is

Dividing throughout by Vinc2 / Zo will give us our Pout / Pin (since Pin is Vinc2 / Zo)

The final step in the conversion to a loss-block is to take 10log10 of this

One last little trick and we are there.


And at last we have our VSWR loss-block (Figure 6)

Figure 6

As a sanity check we should test this to see if the block works for ρ = 1 (100% reflection at plane) and ρ = 0 (No reflection at all)

When ρ = 1, none of Vin gets through the measurement plane and the loss should be infinite
When ρ = 0, all of Vin gets through and the loss should be 0

So testing for ρ = 1


And testing for ρ = 0

But log10[1] = 0, so


As a further step, we can now place this VSWR Loss-Block in front of a perfect match antenna (VSWR = 1:1 over the entire antenna frequency range).

Figure 7

To be continued...

-Tom Mullineaux
Lionheart Southwest